Spread spectrum communication device

ABSTRACT

In a spread spectrum communication system for communicating data by spreading spectrum thereof, using PN codes, on the transmitter side, a first modulated output portion, which is spread-spectrum-modulated, containing no data to be transmitted and a second modulated output portion, which is spread-spectrum-modulated, following the first modulated output portion and containing data to be transmitted are generated; and the second modulated output portion is transmitted with a low electric power and the first modulated output portion with a high electric power. On the receiver side, a correlation-demodulated output is generated by correlation-demodulating a received input; and from the correlation-demodulated output consisting of a first correlation-demodulated output portion corresponding to the first modulated output portion and a second correlation-demodulated output portion corresponding to the second modulated output portion the first correlation-demodulated output portion is detected. Further timing signal serving as a reference for an operation on the receiver side is generated, starting from the first correlation-demodulated output portion.

FIELD OF THE INVENTION

The present invention relates to a method of the spread spectrummodulation communication used in digital data communication and a devicefor realizing same.

BACKGROUND OF THE INVENTION

The spread spectrum communication method is characterized in that it isstrong against disturbance, noise and fading and has properties ofconcealing signals and secrecy and that asynchronous random access canbe effected thereby, as described in "Spread Spectrum Systems" by R. C.Dixon. By this spread spectrum communication method, the initialsynchronization (by which the synchronization between a PN code traincontained in a received signal and a PN code train formed within areceiver is established with a high speed) and the synchronizationholding (by which the synchronization is held stably) are necessary.

As an example of means for giving the initial synchronization describedabove, there is means disclosed in:

Literature 1: JP-B-Sho 64-11178

and on the other hand, as an example of means for giving thesynchronization holding, there is known

Literature 2: Journal of Electronic Communication Society of Japan, 86/4Vol. J69-B

No. 4 pp. 403-405.

The means disclosed in Literature 1 concerning the initialsynchronization is of type, by which a convolver is used as acorrelation demodulator for the receiver As it can be seen from acircuit of the receiver indicated in FIG. 5, in order to have theinitial synchronization of the reference PN code train, time differencebetween the starting point of the reference PN code train and thestarting point of the convolver output is measured; phase differencebetween the PN code train contained in the received signal and thereference PN code train is obtained from this time difference; and thephase of the reference PN code train is regulated so that this phasedifference is removed.

The means disclosed in Literature 2 concerning to the synchronizationholding is similarly of type, by which a convolver is used as acorrelation demodulator for the receiver. As clearly seen from blockdiagrams of a transmitter and a receiver indicated in FIGS. 1A and 1B,respectively, it is characterized in that the phase difference betweenthe PN code train contained in the received signal and the reference PNcode train is converged to zero by initializing the phase of thereference PN code train in the correlation output.

Although the two methods for realizing the initial synchronization andthe synchronization holding described above are useful in principle, inthe case where they were used in a practical wireless propagation path,they had inconveniences that

a. the correlation output cannot be caught because of influences ofnoise and disturbance at effecting the initial synchronization;

b. erroneous operations can be produced by a correlator output otherthan the aimed signal; etc. This is true also for the synchronizationholding operation.

Therefore an example of improvement of the initializing operation isdisclosed in;

Literature 3: JP-A-Hei 1-98338

and an example of improvement of the synchronization holding isdisclosed in;

Literature 4: JP-A-Hei 1-98340.

In Literature 3 stated above, an initial synchronization pattern and apattern for detecting information data start timing are included intransmitted data, and the output of the correlator is led to patternjudging means (i.e. matching filter) in order to assure further theinitial synchronization operation, as shown by the circuit of thereceiver indicated in FIG. 1B, so that influences of noise, etc. arereduced in this way. On the other hand, in Literature 4 concerning thesynchronization holding stated above, it is intended to reduce theinfluences of noise, etc. on the data demodulating operation by reducingthe influences of noise on the synchronization operation by means of adigital filter (i.e. matching filter) and at the same time observingonly the output of the correlator at the neighborhood of the point oftime of the output of the aimed signal by using a window pulse.

However the devices disclosed in Literatures 3 and 4 had problematicalpoints as follows;

a. the circuit is complicated, because the patter judging means, thedigital filter, etc. are necessary;

b. no satisfactory effect can be obtained, in the case where a thresholdvoltage for comparison used at forming two-valued data, because both thepattern judging means and the digital filter perform processing afterthe output of the correlator has been converted into two-valued data;

c. since processing time by the pattern judging means should beincreased in order to increase the effect of the means disclosed inLiterature 3, as the result increase in the time required for theinitial synchronization produced; etc.

OBJECT OF THE INVENTION

Consequently a first object of the present invention is to provide amethod of the spread spectrum modulation communication capable ofrealizing surer catch-up of the correlation output and the signalsynchronization necessary for data demodulation by means of a simplecircuit and a system for realizing same.

Further, in the spread spectrum modulation communication, heretofore, itwas a problem to be solved to provide a modulating-demodulating methodsuitable for this communication method.

For this reason, the applicant of this application has proposed new datamodulating-demodulating methods in his two older patent applications,Nos. Hei 1-29538 and Hei 1-244931. By the data modulating-demodulatingmethod according to the former patent application, as indicated in FIGS.1 and 2 thereof, on the transmitter side a plurality of PN codesdifferent from each other are switched-over according to values ofrelevant data bits to form a spread spectrum code, which is thentransmitted, and on the receiver side an SAW convolver is used for thecorrelator so that data demodulation can be effected without requiringany synchronization between the PN code contained in the received signaland the reference PN code within the receiver. Further the devicedescribed in this patent application, as indicated in FIG. 9 thereof, isso constructed that the reference PN code tracks the PN code containedin the received signal and it is devised to obtain always thecorrelation output. On the other hand the latter patent applicationdescribed above (No. Hei 1-244931) represents an improvement of theformer patent application. In the latter patent application, asindicated in FIG. 1 thereof, a pattern matching circuit and/or a lowpass filter are added to the correlator on the output side thereof toreduce influences of noise and disturbance and in this way the detectionof the presence or absence of the correlation output is effected moresurely so that the tracking method, by which 2 PN codes areswitched-over at detecting disappearance of the correlation output, iseffected more efficiently.

However, in the latter patent application (i.e. patent application No.Hei 1-244931), if performance of the added circuits is raised,in-loop-delay in the tracking loop is increased and thus misstrackingcan take place. Therefore it was difficult to obtain performance over acertain degree. Furthermore, if the in-loop-delay is increased, theperiod of time of the disappearance of the correlation output iselongated. This had bad influences on the circuit generating a thresholdvoltage for comparison, when the output of the correlator is convertedinto two-valued data.

Therefore a second object of the present invention is to provide amethod of the spread spectrum modulation communication capable of makingmore surely the data demodulation possible without using any trackingloop and a system for realizing same.

Still further, in the spread spectrum communication, it is necessary tocontrol suitably the gain of an amplifier before the correlator in thereceiver and an amplifier on the output side of the correlator.

An example of the prior art method for this gain control is indicatedin;

Literature 5: JP-A-Hei 1-109925.

The device disclosed therein is so constructed that, as clearly seenfrom FIG. 1 thereof, the output of the correlator is demodulated; theoutput thus demodulated is compared with a reference voltage; and thegain of a variable gain amplifier connected with the correlator on theoutput side is controlled on the basis of a result of this comparison.However, since this constitutes a negative feed back amplifier so thatthe demodulated output is kept to be constant, in the case where noiseand disturbance are intense and noise, etc. are predominant also in theoutput of the correlator, the noise, etc. act so as to decrease thegain. This was not preferable for the data demodulation.

Therefore a third object of the present invention is to provide a methodof the spread spectrum modulation communication having a gain controlfor assuring the gain necessary for the data demodulation, even in thecase where noise and disturbance are predominant in the output of thecorrelator, and for making the device so as not to judge erroneously forthose noise, etc. to be an aimed signal, in the case where there existsno aimed signal, and a system for realizing same.

SUMMARY OF THE INVENTION

In order to achieve the above first object, a spread spectrumcommunication method according to the present invention is characterizedin that, on the transmitter side, a) a spread-spectrum-modulated outputconsisting of a first modulated output portion, which isspread-spectrum-modulated, containing no data to be transmitted and asecond modulated output portion, which is spread-spectrum-modulated,following said first modulated output portion and containing data to betransmitted is generated; and b) the spread-spectrum-modulated output istransmitted so as to transmit the second modulated output portion with alow electric power and the first modulated output portion with a highelectric power; and on the receiver side, a) a received input iscorrelation-demodulated to generate a correlation-demodulated output; b)from the correlation-demodulated output consisting of a firstcorrelation-demodulated output portion corresponding to the firstmodulated output portion and a second correlation-demodulated outputportion corresponding to the second modulated output portion the firstcorrelation-demodulated output portion is detected; and c) a timingsignal serving as a reference for an operation on the receiver side isgenerated, starting from the first correlation-demodulated outputportion.

In order to achieve the above second object, a spread spectrumcommunication method according to the present invention is characterizedin that, on the transmitter side, a) a first PN code train consisting ofa repetition of first PN codes with a predetermined period and a secondPN code train shifted by a predetermined phase with respect to the firstPN codes is generated; and b) a spread-spectrum-modulated output isgenerated by selecting either one of the first and the second PN codetrain according to each bit of the data to be transmitted so that eachbit of the transmitted data is CPSK-modulated; and on the receiver side,a) a correlation-demodulated output is generated bycorrelation-demodulating a received input with a third PN code trainconsisting of a repetition of second PN codes inverted in time withrespect to the first PN code with the predetermined period; and b) thecorrelation-demodulated output is CPSK-modulated. This CPSK-modulationis effected i) by generating a first time window pulse train consistingof a pulse train with a period, which is a half of the predeterminedperiod, in synchronism with the correlation-demodulated output, a secondtime window pulse train consisting of a pulse train shifted by apredetermined phase with respect to the pulse train stated previously,and a bit duration specifying signal indicating a duration of each bitof the transmitted data; ii) by generating a two-valued pulse outputobtained by converting the correlation-demodulated output with apredetermined threshold; iii) by detecting a difference between a numberof pulses within the first time window and a number of pulses within thesecond time window in the two-valued pulse output in the duration ofeach bit; and iv) by determining a state of the each bit, depending on apolarity of the difference.

In order to achieve the above third object, a spread spectrumcommunication method according to the present invention is characterizedin that, on the transmitter side, a) a spread-spectrum-modulated outputconsisting of a first modulated output portion, which isspread-spectrum-modulated, and a second modulated output portion, whichis spread-spectrum-modulated, following said first modulated outputportion is generated; b) the spread-spectrum-modulated output istransmitted so as to transmit the second modulated output portion with alow electric power and the first modulated output portion with a highelectric power; and on the receiver side, a) a gain in a path fromreception to generation of the correlation-modulated output of thereceived input is set at a low level; b) from thecorrelation-demodulated output consisting of a firstcorrelation-demodulated output portion corresponding to the firstmodulated output portion and a second correlation-demodulated outputportion corresponding to the second modulated output portion an end ofthe first correlation-demodulated output portion is detected; and c) thegain in the path is set at a high level at the detection of the end.

BRIEF DESCRIPTION OF THE DRAWINGS

FIGS. 1A and 1B are block diagrams of a transmitter and a receiver,respectively, used for the spread spectrum modulation communicationaccording to the present invention;

FIG. 2A is a circuit diagram showing in detail a circuit of a controlsignal generating section 16 for determining a transmission format(packet) for the transmitter indicated in FIG. 1A;

FIG. 2B shows waveforms at different parts in the circuit indicated inFIG. 2A;

FIG. 3 is a circuit diagram showing in detail a circuit of a modulatingsection in the transmitter indicated in FIG. 1A;

FIG. 4 shows waveforms at different parts in the circuit of themodulating section indicated in FIG. 3;

FIG. 5 is a circuit diagram showing in detail a circuit of atransmission power control section 15 in the transmitter indicated inFIG. 1A;

FIG. 6 shows waveforms at different parts in the circuit indicated inFIG. 5;

FIG. 7 is a diagram showing the construction of a convolver 22 in thereceiver indicated in FIG. 1B;

FIGS. 8A, 8B and 8C are conceptual diagrams for explaining the operationof the convolver;

FIG. 10 is a circuit diagram showing in detail a circuit of a variablegain amplifier 27 in the receiver indicated in FIG. 1B;

FIG. 11 show a waveform of a gain control signal GC, which is one ofinputs, to the circuit indicated in FIG. 10;

FIG. 12 is a circuit diagram showing in detail a circuit of a detectingcircuit 28 in the receiver indicated in FIG. 1B;

FIG. 13 shows waveforms at different parts in the circuit indicated inFIG. 12;

FIGS. 14A and 14B show waveforms appearing in the circuit indicated inFIG. 12, when CW disturbance takes place;

FIG. 15 is a circuit diagram showing in detail a circuit of a datademodulating section 47 in the receiver indicated in FIG. 1B;

FIG. 16 shows waveforms at different parts in the circuit indicated inFIG. 15;

FIGS. 17A and 17B are diagrams for explaining the necessity of thesynchronization among a received signal, a reference signal and the gateelectrode of the convolver; and

FIGS. 18A, 18B and 18C are diagrams showing another embodiment, in whichburst transmission and reception according to the present invention isapplied to a frequency hopping (FH) type spread spectrum modulationcommunication.

DETAILED DESCRIPTION

Now the present invention will be explained in detail, using anembodiment. Although in the embodiment described below the presentinvention is applied to a direct sequence (DS) spread spectrummodulation communication, the present invention can be applied as wellto a frequency hopping (FH) type spread spectrum modulationcommunication.

FIGS. 1A and 1B are block diagrams of a transmitter 1 and a receiver 2,respectively, used for the spread spectrum modulation communicationaccording to the present invention.

Transmitter 1

At first the transmitter 1 will be explained. The whole constructionconsists of a PN code clock generator 10; a PN code generator 11; amodulating section 13; a high frequency generator 14; a transmissionpower control section 15; a control signal generator 16; a transmittingantenna 17; an interface section 18; and a data clock generator 19.

The clock generator 10 generates a TPNCK clock having a frequency fl forthe transmitter side PN clock as indicated in FIG. 4. The PN codegenerator 11 receiving this clock output TPNCK at a CK terminal thereofgenerates a PN code train having a period T1 at an output terminal OUTthereof.

Here a format of the data transmission for the transmitter 1 will beexplained, referring to FIG. 2B. The transmission format relating to theburst transmission, which is one of the characteristics of the presentinvention, has a packet form including a burst portion containing nodata and a data portion containing data to be transmitted. The controlsignal generating section 16 works so as to determine the duration ofeach packet. An example of that circuit consists of twomonomultivibrators 160 and 162, two analogue switches SW1 and SW2 (eachof the switches being turned-on, when the control input B or C is high),and three resistors Rx, Ry and Rz (Rx<Ry). In this circuit 16, pulses A,B and C can be generated also by a micro-computer. Owing to such acircuit, the control signal generating section 16 generates a controloutput CCS at a control terminal CNT and a start control output SCS at astart terminal START. The voltage of the output CCS at the controlterminal has a positive high value (+V1) in a short period of time fromt1 to t2, i.e. burst period TB (e.g.: 9 μs) and a positive lower value(+V2) in a succeeding period of time from t2 to t3, i.e. data period TD(e.g.: 100 ms), as indicated in FIG. 2B. When this packet period of timeTP is terminated, it has a same negative value (-V0) as before thestart. The other output SCS of the generating section 16 indicates theburst period in the packet and it is high only in that period TB.

Next, a section, which forms a modulating signal, responding to theoutput SCS, will be explained. At first, the data clock generator 19receiving the output SCS at a reset terminal RST generates a data clockDCK having a period Td synchronized with the front edge of the pulse ofthe output SCS (i.e. initialized) at the output terminal OUT, similarlyas indicated in FIG. 2B. This initialization is necessary forsynchronizing the data clock on the receiver side with a detected burstportion in the receiver 2. The period Td determines the duration of eachbit of the transmitted data D. The interface 18, which receives theclock DCK at a clock terminal CK, the output SCS at a terminal B and thetransmitted data D at an input terminal IN, outputs an output I/F, whichis forcedly set at a level corresponding to the high level of the datain the present embodiment during a starting period TS selected so as tobe longer than the burst period TB and includes different bits D1, D2˜DNof the data with the period Td after this period of time TS. This outputI/F acts as a modulating signal. (It is necessary to know to which datalevel the forced setting level during the starting period TScorresponds, when the part corresponding to the burst portion is judged,in the correlation output of the receiver 2. For this reason, the forcedsetting level can be a low level, if it is made correspond to anoperation on the receiver side. Further, if differential coding, whichis a well-known technique, is used, such a forced setting operation isunnecessary, and even if the polarity of the CPSK-demodulated output ofthe receiver, described later, is inverted, correct data reproduction ismade possible by effecting a post differential decoding.)

Next the modulating section 13 effecting the CPSK (code phase shiftkeying), which is one of the characteristics of the present invention,will be described. As it can be seen from FIG. 1A, the modulatingsection 13 is constructed so as to receive a PN code train TPN at an "a"terminal, a clock output TPNCK at a "b" terminal, a modulating outputI/F at a "c" terminal, and a high frequency TRF from the high frequencygenerator 14 at a "d" terminal and to generate a modulated output MODthereof at an "e" terminal.

A detailed circuit of this modulating section 13 will be explained,referring to FIGS. 3 and 4. As indicated in FIG. 3, the modulatingsection 13 is provided with a CPSK modulating section 130 and a BPSK(biphase shift keying) modulating section 132. At first, the CPSKmodulating section 130 is provided with a shift register 1300, twothree-state buffers 1302 and 1304 and an inverter 1306. The buffer 1302receives the PN code train TPN and the state thereof is controlled bythe output I/F. The shift register 300 receives the PN code train TPN atan input terminal IN and a clock TPNCK at a clock terminal CK and asindicated in FIG. 4, it generates a phase shifted PN code train STPNobtained by delaying the input by 1/2 of the period T1, i.e. a phase of180°, at an output terminal OUT. This delay amount (1/2 of T1) in theshift register is preferable, because the greatest phase difference canbe obtained by using it, in the case where the CPSK-modulation iscarried out, and therefore erroneous judgment of the phase can beminimized at demodulation on the receiver side. However it is possiblealso to change this delay amount. Next the state of the buffer receivingthe output STPN is controlled by the output I/F through the inverter1306. Consequently, as indicated in FIG. 4, the buffer 1302 makes the PNcode train TPN pass through, as long as the output I/F is high, whilethe buffer 1304 makes the phase shifted PN code train STPN pass through,as long as the output I/F is low. In this way a CPSK modulated outputCPSKMOD is formed. At this modulation, since the frequency f1 of the PNcode clock TPNCK is selected so as to by very high with respect to thefrequency f2 of the data clock DCK, the spectrum of the I/F output isspread in the modulated output CPSKMOD (spread spectrum modulation).

The succeeding BPSK modulating section 132 consists of a D-type flipflop(F/F) 1320, two resistors R1 and R2, and a double balanced modulator1322. More in detail, the D-type F/F 1320 receives a modulated outputCPSKMOD at a D terminal, a clock TPNCK at a CK terminal, and outputs anormal phase output at a Q terminal and an inverted phase output at a Q*terminal (* indicating inversion) to differential-drive the modulator1322 receiving the high frequency TRF by these two outputs. As indicatedin the figure, the modulator 1322 is provided with two transformers T1and T2 and a diode bridge DB and generates an output calculated bymultiplying the high frequency by the output CPSKMOD as a doublebalanced modulated output and therefore the final modulated output MODof the modulating section 13.

Next the transmission power control section 15 transmitting thismodulated output will be explained. This control section is a partrelating to the burst transmission, which is one of the characteristicsof the present invention, and it receives the modulated output MOD at aninput terminal IN1 and the output CCS of the control signal generatingsection 16 at the other input terminal IN2 to supply a power controltype transmission output TPC from the output OUT to the antenna 17. Asindicated in FIG. 5, the detailed circuit thereof consists of twotransformers T3 and T4, two diodes D1 and D2 and one resistor R3. Oneend of the primary winding of the transformer T3 is coupled with themodulated output MOD and the other end thereof is grounded, while thecenter tap of the secondary winding thereof is coupled with the controloutput CCS through the resistor R3 and further the two ends thereof areconnected with the two ends of the primary winding of the othertransformer T4 through the diodes D1 and D2, respectively (the centertap thereof being grounded). One end of the secondary winding of thistransformer T4 is grounded and the other end thereof is connected withthe output terminal OUT. This circuit functions so as toamplitude-modulate the modulated output MOD with the control output CCS.Since the high frequency resistance of the diodes D1 and D2 decreases,as the voltage of the control output CCS increases positively, an outputhaving a great amplitude is generated at the OUT terminal.

Consequently, as indicated in the diagrams representing waveforms inFIG. 6, because of the high voltage +V1 of the control output CCS duringthe burst period TB, the transmission output TPC has a great amplitude,which is Vb (e.g. the peak electric power of the burst part being 1 mW).On the contrary, during the data period TD, since the control output CCShas the low voltage +V2, the transmission output TPC has a smallamplitude Vd (e.g. the average electric power of the data part being 10μW). Outside of the packet period TP the transmission output TPC isextremely small or zero.

In the transmitter 1 as described above, it is necessary to select theduration of the burst period, relating it with the processing time ofthe correlator used in the receiver 2. Denoting the processing time ofthe correlator by T, it is preferable to select the duration of theburst period so that it satisfies;

    TB≧1.5T

In the case where an SAW convolver is used for the correlator, theprocessing time T is determined by the gate length thereof.

Receiver 2

Now the receiver 2 will be explained.

As indicated in FIG. 1B, the receiver 2 consists of RF/IF (highfrequency amplification, frequency conversion, intermediate frequencyamplification) section 21; an SAW convolver 22 acting as the correlator;a reference signal generating section including a receiver side PN codeclock generator 23, a receiver side PN code generator 24, a highfrequency generator 25 and a double balanced modulator 26; a variablegain amplifier 27; a detecting circuit 28; and a data demodulatingsection 29. The circuits 22 to 28 constitute correlation demodulatingmeans.

More in detail, the clock generator 23 generates a clock output RPNCKhaving the same clock frequency fl as the clock for the transmitter sidePN code and the generator 24 receiving it at a CK terminal outputs a PNcode train RPN inverted in time with respect to the transmitter side PNcode. The modulator 26 receiving this output is connected so as toreceive a high frequency output RPG having the central frequency of afrequency converted received signal FCS from the high frequencygenerator 25 as another input. The double balanced modulator 26modulates a high frequency RPG with the PN code train RPN to form areference signal RS used by the convolver 22.

The convolver 22 receiving the received signal FCS and the referencesignal RS at an IN1 terminal and an IN2 terminal, respectively,generates a convolution output CONV at an OUT terminal thereof. More indetail, the convolver 22 can be an SAW convolver 22A having theconstruction indicated in FIG. 7 (221 and 222 being interdigitaltransducers IDT; 223 being a gate electrode; 224 being a zinc-oxide(ZnO) layer; 215 being a silicon oxide (SiO₂) layer; 226 being a silicon(Si) layer; and 227 being an ohmic electrode). At this time, denoting anideal signal FCS received at the IN1 terminal, containing noise, by afunction s(t) and similarly an ideal reference signal RS received at theIN2 terminal by a function r(t), the convolution output CONV at thistime, i.e. a function c(t), is given by a following equation;

    c (t)=∫τs (τ)·Γ(2t-τ-T) dτ

where T represents the processing time of the convolver, as describedpreviously. Here supposing that the period T1 of the PN code is 9 μsecand the processing time T is also 9 μsec, since the PN codes are inaccordance with each other at t=0, t=4.5 μsec and t=9 μsec, as indicatedconceptually in FIG. 8A, 8B and 8C, a correlation output with a periodTcsp, i.e. great for every 4.5 μsec, can be obtained at the convolutionoutput c(t), as indicated in FIG. 9. This is true in the case where s(t)and r(t) are continuous signals. On the contrary, in the case of adiscontinuous signal as the burst portion according to the presentinvention, in order to obtain a correlation output, which is as great asthat obtained for the continuous signal, it is necessary that the burstperiod TB is longer than 1.5 times of the processing time T of theconvolver. (It has been already described that TB≧1.5T is preferable).

Next the variable gain amplifier 27 receiving the convolution outputCONV, as described above, will be explained, referring to FIGS. 10 and11. This variable gain amplifier relates to one of the characteristicsof the present invention. As indicated in FIG. 10, this variable gainamplifier 27 is provided with an RF amplifier 270 connected in seriesbetween the IN terminal and the OUT terminal; a capacitor Cl; a variableattenuator 272; a capacitor C2; and an RF amplifier 274. In the variableattenuator 272, as indicated in the figure, a point X is groundedthrough a forward direction diode D3 and a capacitor C3; a point Y isgrounded through a reverse direction diode D4 and a capacitor C4 as wellas a resistor R4 connected in parallel with them; and the point X isconnected with the point Y through a capacitor C5 and a forwarddirection diode D5. Further the point X is connected with a fixed DCpower supply El through a resistor R5. Furthermore a point Z isconnected with a point W through a coil I and this point W is connectedwith a GAIN terminal through a reverse direction diode D6, a resistor R6and an inverter 276. In addition the point W is grounded through areverse direction diode D7, a resistor R7 and a fixed DC power supplyE2.

In this circuit indicated in the figure, the variable attenuator 272works so as to decrease the attenuation (increasing the gain of theamplifier 27), as the potential at the point W rises positively.Consequently, when it receives a gain control output GC (describedlater), as indicted in FIG. 11, at the GAIN terminal, during a periodwhere the level is high, the drive path in the attenuator 272 isE2→R7→D7 and lowers the voltage at the point W to set a low gain G_(L).On the contrary, during a period where the level is low, the drive pathis 276→R6→D6 and raises the potential at the point W to set a high gainG_(H). The magnitude of the gain thus set is determined by selecting theoutput voltage of the inverter 276, the voltage of the power supply E2and resistances of the resistors R6 and R7. As the result, this variablegain amplifier 27 generates an output ACONV amplified with a variablegain of the output CONV at the OUT terminal.

Next the detecting circuit 28 receiving this output ACONV at the INterminal will be explained, referring to FIGS. 12, 13, 14A and 14B. Thedetecting circuit 28 relates to the characteristics of the gain controlaccording to the present invention and is provided, as indicated indetail in FIG. 12, with a double balanced modulator 280 receiving theoutput ACONV of the amplifier 27 at the two inputs; a band pass filter282 receiving an output DBM thereof; an amplifier 284 receiving anoutput BPFO of this band pass filter; and a double balanced modulator286 receiving the output of the amplifier at the two inputs thereof forthe purpose of rectification. The waveform within this detecting circuitis indicted in FIG. 13 in an ideal state and in FIGS. 14A and 14B, inthe case where CW disturbing signals are mixed therein in the antenna20.

FIG. 13 shows the output ACONV in an ideal state, which is obtained byenlarging the waveform indicated in FIG. 9. It has a frequency component(which, in this case, contains no disturbing component) having a centralfrequency f₀ as indicated in FIG. 14B. Further T_(CSW) is equal to thedriving clock frequency for the PN code. Receiving this output, themodulator 280 generates the output DBM by squaring it. This output has aspectrum containing a doubled frequency component (the central frequencybeing 2f₀), as indicated in FIG. 14B. As it can be seen from the outputDBM' indicated in FIG. 14B, the correlation components between the CWdisturbing signals and the reference signal RS are concentrated in theneighborhood of the direct current in the spectrum. On the contrary, thecorrelation components between the aimed signal component and thereference signal RS is distributed over a wide frequency band in thespectrum. The pass band f_(L) to f_(H) of the filter 282 is so selectedthat the lower frequency limit f_(L) is higher than the CW disturbingcomponents in the neighborhood of DC in the output DBM, and that theupper frequency limit f_(H) is equal to the upper limit of the primaryaimed component in the output DBM'. The output BPFO of this band passfilter 282 is rectified by the succeeding modulator 286 to produce thefinal detected output ADCONV indicated in FIG. 13.

In the case where there is CW disturbance, the convolution output ADCONVis represented by ADCONV' indicated in FIGS. 14A and 14B and the outputof the modulator 280 at this time is DBM'. At this time the majority ofthe CW disturbing components is eliminated from the output BPFO' of theband pass filter. (In the case where no special measures for suppressingthe CW disturbance is required, the detecting circuit 28 may be a diode,which effects merely envelope detection.)

At last, the data demodulating section 29 effecting data demodulationfrom this detected output ADCONV will be explained, referring to FIGS.15 and 16. The data demodulating section 29 receives the clock outputRPNCK at an "a" terminal and the detected output ADCONV at a "b"terminal to output the demodulated data at a "c" terminal and the gaincontrol signal GC at a "d" terminal. In detail, as indicated in FIG. 15,this circuit is composed of a burst detecting circuit 290 effectingburst detection, a timing signal generating circuit 292 generatingtiming signals used for various sorts of demodulations in synchronismwith the burst thus detected and a CPSK demodulating circuit 294,roughly divided. FIG. 16 shows various sorts of waveforms in thesecircuits.

At first, the burst detecting circuit 290 is a part relating to thecharacteristics of the burst transmission and reception according to thepresent invention and includes a comparator 2900, which compares theconvolution output ADCONV, which has been detected, (the time axis inFIG. 16 being compressed with respect to those used in FIGS. 13 and witha fixed threshold voltage E3 (indicated also in FIG. an AND gate 2904and a timer 2902. The height of the threshold E3 used in the comparator2900 is selected so as to be smaller than the convolution outputproduced by the burst portion, but greater than the convolution outputproduced by the data portion, i.e. P_(B). In FIG. 16, an ideal outputADCONV containing no noise and an output ADCONV" containing noise areindicated. In the following description, explanation will be made forthe latter output ADCONV" containing noise.

The comparator responding to the output ADCONV" generates an output C1containing only pulses corresponding to a pulse P_(B). The AND gate 2904receiving this output C1 is constructed so as to receive an output TM1of the Q* terminal of the timer 2902 as the other input. This timermakes the timer output TM1 usually high and low during a certain timeTh, when a trigger input is received by a trigger terminal TRIG. Thistime Th is selected so as to be longer than the packet period TP.Consequently the gate output AG of the AND gate generates a pulsecorresponding to the pulse P_(B). At this time the timer output TM1 ismade low during the time Th. During this time, Th, i.e. at least duringthis packet

period TP, the AND gate obstructs pulses P1˜P28 . . . succeeding thepulse P_(B). In this way it is secured that only the burst portion actsas the time reference and the other part is not used as the timereference. The timer output TM1 is the gain control output GC indicatedin FIG. 11.

The following timing signal generating circuit 292 is provided with agate pulse generator 2920 and a shift register 2922 for forming two timewindow pulse trains; as well as a data clock generator 2924 and a D-typeF/F 2926 for generating a signal indicating a duration of each data bitand a sampling clock for the demodulated data. The generator 2920receives the gate output AG at a reset terminal RST and the clock outputRPNCK at a CK terminal thereof to generate a pulse after a lapse of timeT_(CSP) measured from the rise of the gate output AG as the output GPand to generate repeatedly pulses thereafter with a period of T_(CSP).These gate pulses, which are the output GP, function as "1" side timewindow pulses. Next, the shift register 2922 receiving the output GP atthe IN terminal and the clock RPNCK at the CK terminal delays the gatepulse by a phase of 180° (which corresponds to a phase delay of 180° atthe shift register indicated in FIG. 3 and to 1/2 of the delay time bythe shift register 1300 as the delay time, because the time axis for theconvolver output is compressed to 1/2 with respect to the timer axis forthe input thereof). Consequently it generates a phase shift gate pulseoutput SGP at the OUT terminal. Pulses of this phase shift gate pulseoutput SGP function as "0" side time window pulses. Further the dataclock generator 2924 receiving the gate output AG at the RST terminaland the clock RPNCK at the CK terminal outputs data clock outputs DCrepeatedly at the OUT terminal with a period Td (the data clock periodindicated in FIG. 2) after the rise of the pulse in the output AG. Thisoutput DC functions as a sampling clock for the demodulated data. TheD-type F/F 2926 receiving this output at the D terminal and the clockoutput RPNCK at the CK terminal generates a pulse obtained by delayingthe output DC by one period of the clock output RPNCK as an output FO.This output FO is used for initializing the circuit for thedemodulation.

The synchronization by using the burst described above has an advantageas follows. Firstly, since certainty of the synchronization can beraised at need by increasing the transmission power for the burstportion with respect to the data portion, sure synchronization is madepossible even in an environment, where noise/disturbance is intense.Further, in this way, it is possible to raise the disturbance excludingproperty up to the limit property of the data demodulation by thereception of the data portion. Secondly, since the burst period TB canbe selected so as to be about as long as the processing time of the usedcorrelator, it can be very short. For example, in the case of aconvolver having a processing time of 9 μsec, the burst period can be 9to 14.5 μsec. Consequently, in the case where packet transmission iseffected, the transmission efficiency can be increased by shortening thepart of previous processing and therefore it is suitable for a highspeed transmission. Thirdly, since the burst portion can be extremelyshort, together with the spread spectrum modulation, although electricpower is high, disturbance given to other communication is small.Fourthly, although the peak electric power of the burst portion is high,since the ratio of the transmission time therefor to that required forthe data portion is small and thus increase in the average electricpower is slight, low electric power consumption can be realized.Fifthly, since the peak electric power of the burst portion is high,carrier sense (to verify whether another station is effectingtransmission or not, by carrying out previously reception beforetransmission, at constructing a wireless network, where a plurality ofstations effect random access) is made easier.

Next the CPSK demodulating circuit 294 will be explained. This is one ofthe characteristics of the present invention and provided with acomparator 2940 receiving the detected output ADCONV (in the presentexplanation ADCONV") and comparing it with a threshold voltage DAO forconverting input data into two-valued data (explained later); two ANDgates 2942 and 2944; two timers 2946 and 2948; and a counter 2950.Further an OR gate 2952, a binary counter 2954 and a D/A converter 2956constitute a sweep type threshold voltage control circuit for convertingdata into a two-valued signal, which varies the threshold voltage DAO.Owing to this circuit portion the threshold voltage has valuesincreasing stepwise, as indicated in chain-dotted line on the outputADCONV" in FIG. 16

The comparator 2940 for converting data into a two-valued signal, whichcompares the input with such a threshold voltage DAO, generates twopulses due to noise components between two pulses P1 and P2 in additionto pulses corresponding to pulses P1, P2, P3, P6 and P8 in the data bitD1 section. In the data bit D2 section it outputs P13, P14, P15, P17,P20 and P24 as well as three pulses due to noise between P13 and P24.The AND gate 2942 receiving the output C2 of this comparator receivesthe "1" side time window pulse train GP as another input. On the otherhand the AND gate receives that output C2 and the "0" side time windowpulse train SGP and consequently the gates 2942 and 2944 act so as tosend pulses of generation timings corresponding to data "1" and "0" toTRIG terminals of the succeeding timers 2946 and 2948, respectively.When these timers receive pulse inputs at the respective TRIG terminals,they keep the potentials at the respective Q terminals at a high levelfor a certain period of time. By the respective trigger inputs theycontinue to keep them further at the high level for the certain periodof time. Further the length of that certain period of time is selectedso as to be greater than the pulse width of each time window pulse andsmaller than the period of the time window pulse so that only one pulseis outputted within each time window. As the result the timer outputsTM2 and TM3 of the different timers are as indicated in the figure.

Now the sweep type threshold voltage control circuit for converting datainto a two-valued signal will be explained. As indicated in FIG. 6, theOR gate 2952 receiving the timer outputs TM2 and TM3 at the two inputsthereof generates the gate output OG including all the pulses withinthese timer outputs. The binary counter 2954 of N2 bits receiving thisoutput OG at a CK terminal and the output FO at a clear terminal CLRgenerates an output BC obtained by counting the pulses. This countoutput BC is converted into an analogue value by the succeeding D/Aconverter 2956 to form the threshold voltage DAO. The aspect, with whichthis voltage DAO increases stepwise with increasing count, is asindicated in the figure. When a pulse is produced in the output FO atthe end of each data bit section, the counter 2954 is cleared. As theresult, the threshold voltage DAO is returned to zero volt.

By such a sweep of the threshold voltage from zero volt to the peakvalue of the correlation spike during the data period, since thedifferent parts during each of the data bit periods are examined byusing the different thresholds, an effect can be obtained that asubstantially constant property can be obtained, independently ofreception conditions. Consequently, comparing it with the case wherethis threshold is set at a certain fixed value, the judgment of theoptimum level becomes unnecessary and even in the case where disturbanceand noise are intense, it can be used in practice. Further, since thethreshold voltage is swept stepwise only in one direction (in thisexample of explanation, in the positive direction), comparing it withthe following type threshold control, there is no risk of oscillationand therefore it is suitable for high speed response. Furthermore, sincethe operation is independent for every data bit, it has an advantagethat, even if the operation is unsuitable for a certain bit, this has noinfluences on the operation for the succeeding bits.

Next the counter 2950 of N1 bits and the D-type F/F 2958, which judgethe state of each data bit on the basis of the numbers of pulses withinthe timer outputs TM2 and TM3, will be explained. In this part, thepolarity of the difference between these numbers of pulses is used forjudging the state "0" or "1" of the data bit (the absolute value of thedifference having nothing to do). The counter 2950 receives the timeroutput TM2 at the up-count input U and the timer output TM3 at the downcount input D and in addition fixed input data, i.e. 2^(N1-1) (if N1=8,this being 128, i.e. in the binary representation 10000000) at the INterminal and the output FO at the load terminal LD so as to output themost significant bit MSB at the OUT terminal. Therefore, in the casewhere the difference obtained by subtracting the number of pulses withinthe timer output TM3 from the number of pulses within the timer outputTM2 in each data bit is positive, the most significant bit MSB is thestate "1" and the data bit at that time is interpreted to be "1" . Onthe contrary, in the case where the difference is negative, the data bitis interpreted to be "0". This MSB output is sampled by the output DC inthe succeeding D-type F/F to produce the demodulated data. The output FObecomes high immediately after the sampling clock DC so that fixed inputdata are read into the counter 2950. The operation described above isrepeated for the succeeding bit.

Further, in the example described above, the fixed input data can bechanged from the value stated above of 2^(N1-1) to 2^(n1) -1 or it canbe changed further for the last value to another in a region, whereefficient use of the count range is not prevented.

The CPSK modulation-demodulation by the CPSK demodulation explainedabove and the CPSK modulation stated above has an advantage as describedbelow. Firstly, since the correlation pulse is selectively extracted byusing two kinds of time window pulses, noise or disturbance, which canexist in a period of time outside of the time windows, is not caught andthus the effect of suppressing noise/disturbance is remarkable. Further,owing to the type of processing, where the correlation output doesn'tdisappear in principle, the noise/disturbance suppressing effect can beefficient by adding correlation outputs (adding operation by means of acounter). Secondly, owing to the type of processing, where the polarityof the difference between the numbers of the two kinds of correlationpulses, which have passed through those two time windows, is madecorrespond to the state of the demodulated data, the data demodulationis substantially independent of the absolute value of the correlationpulses and hardly influenced by circuits outside of the demodulationsection. Therefore the stability of the data demodulation is increased.Further, since any tracking loop as used heretofore is not used, thereis no fear of oscillation.

Now the operation of the variable gain amplifier 27 indicated in FIG.10, receiving the gain control signal GC (indicated in FIGS. 16 and 11)of the demodulating section 29, which has been explained just above,will be explained. At it can be seen from FIG. 16, the gain controlsignal GC is high and sets the amplifier 27 at a low gain G_(L) in theperiod other than the packet period TP and in the burst period TB of thepacket, i.e. in the period, where the burst portion is waited, so thatnoise, etc. are not detected, erroneously judging them to be a burstportion. On the other hand, the signal GC is low and sets the amplifier27 at a high gain G_(H) during a period of time of Th after thedetection of the burst portion, i.e. at least in the data period TD. Inthis way the data portion having a low transmission power (e.g. 1/100 ofelectric power for the burst portion: 10 μW/1 mW) can be surelydetected, even in the case where disturbance, etc. are intense. For sucha gain control, it is not necessary to effect any negative feedbackcontrol of the output of the correlator as required by the prior arttechnique.

Such a gain control is desirable specifically, in the case where adetecting circuit as indicated in FIG. 12 is used. This is because thedetecting circuit 28 is useful against CW disturbance but noise iscontained abundantly, when S/N of the input of the receiver is small(i.e. in the case where noise is intense), and at this time, when thegain control stated above is used, it is possible to prevent erroneousoperation due to noise in the output of the detecting circuit.

In the example indicated in the fire, the amplifier 27, which is on theoutput side of the convolver 22, is controlled by the gain controlsignal GC. However, instead thereof, the RF/IF section 21 can becontrolled in the gain, as indicated by a broken line 30 in FIG. 1B.

Although in the receiver 2 explained above, an example, in which aconvolver is used for the correlator, has been explained, a matchedfilter having fixed sign or a digital correlator can be used for thecorrelator. The matched filter, which is the former, is one, in whichthe pattern of the interdigital electrodes is made in accordance withthe pattern of the used PN code; the frequencies of the input and theoutput thereof are equal to each other; and the time axis doesn't vary.The digital correlator, which is the latter, is one, in which theoperation of the matched filter is constructed by using digitalcircuits. Although operations are effected in the base band, it canrealize an operation equivalent to that of the matched filter.

Application of the burst transmission/reception to other systems

The burst transmission/reception explained above can be applied also tothe method, by which the phase of the PN code of the reference signal issynchronized, as described in Literature 1 (JP-B-Sho 64-11178). At thistime the timing signal obtained from the received burst signal can beused for the phase synchronization therefor.

The reason why such a phase synchronization is necessary will beexplained, referring to FIGS. 17A and 17B. In the case where a convolveris used for the correlator, when the period T1 of the PN code isapproximately equal to the smallest width of the data, i.e. the dataclock period Td, the phase synchronization between the PN code of thereceived signal and the PN code of the reference signal on the receiverside is necessary. (In the case where T1<Td, such a phasesynchronization is unnecessary.) FIG. 17A shows the state of a correctphase synchronization among the received signal, the reference signal,and the gate electrode of the convolver. (+PN indicates a PN codecorresponding to data "1", while -PN indicates a PN code correspondingto data "0".) At this time the greatest correlation output can beobtained. On the other hand, FIG. 17B indicates a case where the phasesof the two signals are in accordance with each other, but they aredeviated from the gate electrode. In this case the correlation outputdecreases and therefore this is not preferable. This is the reason whythe phase synchronization is necessary.

Further, the burst transmission/reception according to the presentinvention can be used for the spread spectrum modulation communicationnot only by the direct sequence (DS) system but also by the frequencyhopping (FH) system. By this FH system, as indicated in FIG. 18A, dataare transmitted while changing over the transmission frequency with ahigh speed as f₁, f₂, f₃ . . . On the receiver side, the data arereceived by changing over the reception frequency in synchronism withthe change-over of the transmission frequency. Also by this system thesynchronization control of the received signal is inevitable.Consequently, as indicated in FIG. 18A, transmission of the burstportion BP by the DS system is effected at the beginning of atransmission packet and a frequency hopping signal FH is transmitted byusing this burst transmission as the time reference. On the other hand,on the receiver side, as indicated in FIG. 8B, the hopping frequency ofthe frequency synthesizer within the receiver is controlled by using thecorrelation pulse output PP obtained by receiving the burst portion BPto synchronize the reception frequency with the transmission frequency.FIG. 18C is a block diagram of a receiver 4 effecting such an operation.(The parts analogous to those indicated in the receiver 2 in FIG. 1B areindicated by same reference numerals, to which "a" is added.) As it isseen from the figure, it is so constructed that a correlation pulse PPoutputted by the comparator 2900a as a comparison result with thethreshold voltage E3a is inputted to a start terminal START of thefrequency synthesizer 44. When this is inputted therein, the synthesizer44 begins the frequency hopping, starting from a predetermined frequency(f₁ in this example). The mixer 45, the IF amplifier 46 and the datademodulating section 47 receiving the output of this synthesizer and theoutput of the RF amplifier 40 are identical to those used by the priorart technique.

Further the synchronization by the burst according to the presentinvention can be applied to all other sorts of synchronizing operation(including synchronization such as synchronization of the time windowsfor the CPSK-demodulation described previously) in a receiver for thespread spectrum modulation communication.

According to the present invention explained above in detail, variouseffects such as remarkable effect of suppressing noise/disturbance, highspeed transmission, small disturbance to other communication, lowelectric power consumption, possibility of carrier sense, etc. can beobtained by the burst transmission/reception. Further, by the gaincontrol and the CPSK modulation-demodulation according to the presentinvention, an effect can be obtained that the system is strong againstnoise/disturbance and a high stability is obtained.

What is claimed is:
 1. A spread spectrum communication method forcommunicating data by spreading the spectrum thereof, comprising:A) onthe transmitter side:a) a step of generating a spread-spectrum-modulatedoutput including a first modulated output portion which isspread-spectrum-modulated and contains no data to be transmitted and asecond modulated output portion which is spread-spectrum-modulated,follows said first modulated output portion, and contains data to betransmitted; and b) a step of transmitting saidspread-spectrum-modulated output so as to transmit said second modulatedoutput portion with a low electric power and said first modulated outputportion with a high electric power; and B) on the receiver side:a) astep of correlation-demodulating a received input to generate acorrelation-demodulated output which includes a firstcorrelation-demodulated output portion corresponding to said firstmodulated output portion and a second correlation-demodulated outputportion corresponding to said second modulated output portion; b) a stepof detecting from said correlation-demodulated output said firstcorrelation-demodulated output portion, wherein only a portion of saidcorrelation-demodulated output exceeding a predetermined threshold isdetected as said first correlation-demodulated output portion; and c) astep of generating a timing signal serving as a reference for anoperation on the receiver side, starting from said firstcorrelation-demodulated output portion.
 2. A method according to claim1, wherein said operation on the receiver side includes the step ofusing said timing signal for demodulating said transmitted data fromsaid second modulated output portion.
 3. A method according to claim 1,wherein a period of said first modulated output portion is shorter thana period of said second modulated output portion.
 4. A method accordingto claim 1, wherein said spread spectrum modulation is of directsequence type.
 5. A method according to claim 4 wherein said directsequence type spread spectrum modulation is CPSK modulation.
 6. A methodaccording to claim 1, wherein said spread spectrum modulation is offrequency hopping type.
 7. A spread spectrum communication device forcommunicating data by spreading the spectrum thereof, comprising:A) atransmitter which includes:a) period specifying means for determining apacket period which includes a burst period containing no data to betransmitted and a data period following said burst period and containingdata to be transmitted; b) PN code train generating means for generatinga PN code train which includes a repetition of first PN codes with apredetermined period; c) modulating signal generating means forgenerating a modulating signal which includes said data to betransmitted during said data period and includes no data during saidburst period; d) spread-spectrum-modulating means for generating aspread-spectrum-modulated output by spread-spectrum-modulating said PNcode train with said modulating signal; e) transmission power specifyingmeans for generating a transmission power specifying signal whichspecifies a high transmission power during said burst period and a lowtransmission power during said data period; and f) transmission powercontrol means for transmitting said spread-spectrum-modulated outputwith an electric power corresponding to said transmission powerspecifying signal; and B) a receiver which includes:a) means forobtaining a frequency-converted output by frequency-converting areceived input; b) correlator means for generating a correlation outputfrom said frequency-converted output by using a PN code train whichincludes a repetition of second PN codes inverted in time with respectto said first PN code; c) burst detecting means for comparing saidcorrelation output with a predetermined threshold and for generating atiming signal when said correlation output is greater than saidpredetermined threshold, said predetermined threshold being lower than alevel of said correlation output during said burst period but higherthan the level of said correlation output during said data period; andd) data demodulating means for demodulating said correlation output byusing said timing signal as a time reference.
 8. A device according toclaim 7 wherein said burst period is shorter than said data period.
 9. Aspread spectrum communication device for communicating data by spreadingthe spectrum thereof, comprising:A) a transmitter which includes:a)period specifying means for determining a packet period which includes aburst period containing no data to be transmitted and a data periodfollowing said burst period and containing data to be transmitted; b) PNcode train generating means for generating a PN code train whichincludes a repetition of first PN codes with a predetermined period; c)modulating signal generating means for generating a modulating signalwhich includes said data to be transmitted during said data period andincludes no data during said burst period; d) spread-spectrum-modulatingmeans for generating a spread-spectrum-modulated output byspread-spectrum-modulating said PN code train with said modulatingsignal; e) transmission power specifying means for generating atransmission power specifying signal which specifies a high transmissionpower during said burst period and a low transmission power during saiddata period; and f) transmission power control means for transmittingsaid spread-spectrum-modulated output with an electric powercorresponding to said transmission power specifying signal; and B) areceiver which includes:a) means for obtaining a frequency-convertedoutput by frequency-converting a received input; b) correlator means forgenerating a correlation output from said frequency-converted output byusing a PN code train which includes a repetition of second PN codesinverted in time with respect to said first PN code; c) burst detectingmeans for comparing said correlation output with a predeterminedthreshold and for generating a timing signal when said correlationoutput is greater than said predetermined threshold, said predeterminedthreshold being lower than a level of said correlation output duringsaid burst period but higher than the level of said correlation outputduring said data period; and d) data demodulating means for demodulatingsaid correlation output by using said timing signal as a timereferencewherein said transmission power control means includes: a) afirst transformer receiving said spread-spectrum-modulated output by aprimary winding thereof and said transmission power specifying signal bya center tap of a secondary winding thereof; and b) a second transformerincluding a primary winding with two extremities which are connectedwith two extremities of said second winding of said first transformerthrough two diodes, respectively, and whose center tap is grounded, anda secondary winding for generating an output.
 10. A device according toclaim 7, wherein said spread-spectrum-modulating means effects directsequence type spread spectrum modulation.
 11. A device according toclaim 10 wherein said direct sequence type spread-spectrum-modulation isCPSK modulation.
 12. A device according to claim 8 wherein saidspread-spectrum-modulating means effects frequency hopping type spreadspectrum modulation.